Switched mode power supplies are well known in the art.
For example, FIG. 1 shows the general architecture of a DC-DC boost converter, which receives a DC input voltage VBAT and provides a DC output voltage VOUT, which is greater than the input voltage VBAT. For example, the input voltage VBAT may be provided via two input terminals 102a and 102b, wherein the negative input terminal 102b is connected to ground GND, and the output voltage VOUT may be provided via two output terminals 104a and 104b, wherein the negative output terminal 104b is connected to ground GND.
For example, the input voltage VBAT may be provided by any DC power source, such as a battery or a rectified AC voltage, and the output voltage VOUT may be used to power a load. Usually, the output voltage VOUT is a design parameter and is set in advance.
Conventional boost DC-DC converters typically comprise an inductive energy storage element L, e.g. an inductor, and a “free-wheeling” diode rectifier D, which are connected in series between the positive input terminal 102a and the positive output terminal 104a. The diode rectifier D conducts current only in the direction of the load, i.e. the anode is connected to the inductive element L and the cathode is connected to the positive output terminal 104a, and is considered free-wheeling because its operation cannot be controlled independent of the voltages at its anode and cathode. A boost converter usually comprises also an output capacitor COUT connected between the output terminals 104a and 104b, which stabilizes the output voltage VOUT.
In the example considered, the boost converter comprises moreover a switch SW, such as a metal-oxide-semiconductor field-effect transistor (MOSFET) of the n-type. Specifically, this switch SW is configured to selectively connect the intermediate point between the inductive element L and the diode D to ground GND as a function of a control signal DL, i.e. the switch SW is connected between this intermediate point and ground GND.
Accordingly, when the switch SW is closed, a current IBAT flows from the positive input terminal 102a and through the inductor L to ground GND, and the inductor L stores energy by generating a magnetic field. Conversely, when the switch SW is opened, current will be reduced as the impedance is higher and the previously created magnetic field will generate a current flow through the diode D and towards the positive output terminal 104a, i.e. the capacitor COUT and the load. Those of skill in the art will appreciate that the diode D may also be replaced with a second switch, which is driven with a signal being substantially complementary to the signal DL, e.g. a signal corresponding to an inverted version of the signal DL.
In this context, a voltage mode control circuitry is often used to control the operation of the boost converter. Specifically, this control circuit performs a closed loop control operation by monitoring the output voltage VOUT and controlling the switching of the switch SW in order to regulate the output voltage VOUT to a desired value.
For example, the circuit may comprise a voltage divider comprising at least two resistors R1 and R2, which are connected in series between the output terminals 104a and 104b, i.e. in parallel with the output capacitor COUT. Accordingly, based on the well-known operation of a voltage divider, the intermediate point between the resistors R1 and R2 provides a voltage VFB being indicative for the output voltage VOUT.
In the example considered, this voltage VFB is provided to a control unit 20, which controls the switching operation of the switch SW as a function of the feedback voltage VFB.
Generally, different control schemes may be implemented within the control unit 20 for controlling the switch SW as a function of the feedback voltage VFB, which are well known to those skilled in the art.
For example, the control circuit 20 may comprise a driver circuit 22 and a detection circuit 24.
For example, the detection circuit 24 may be an error amplifier and the driver circuit 22 may be a pulse-width-modulation (PWM) driver circuit. In this case, the error amplifier 24 is configured to amplify the difference between the voltage VFB with a reference voltage VREF. For example, the reference voltage VREF may be derived from the input or output voltage via a voltage regulator or a band-gap reference. The signal at the output of the error amplifier 24 may then be provided to the driver circuit 22, which varies, i.e. increase and/or decreases, the duty cycle of the PWM driver signal DL such that the error signal at the output of the amplifier 24 is substantially zero.
The detection circuit 24 may also comprise only a simple comparator, which compares the voltage VFB with the reference voltage VREF. In this case, the driver circuit 24, may generate a pulsed driver signal DL until the signal at the output of the comparator 24 indicates that the feedback voltage VFB has reached the reference voltage VREF.
Those of skill in the art will appreciate that also other driving schemes may be used, such as frequency modulation or quasi resonant driving. Moreover, in case of small output loads, the driver circuit 24 may also be deactivated for given time periods and the switch SW may remain opened, usually called “burst mode”.
The inventors have observed that independently from the specific implementation of the control unit 20, the (medium) output voltage VOUT will be:
                              V          OUT                =                              (                          1              +                                                R                  1                                                  R                  2                                                      )                    ⁢                      V            REF                                              (        1        )            
However, at no load condition, i.e. with the load disconnected or switched off, and supposing an ideal efficiency of 100%, there will still be a current consumption IRES by the voltage divider R1/R2 and accordingly a current flow IBAT from the input 102a, which corresponds to:
                              I          BAT                =                                            I              RES                                      V              BAT                                ⁢                      V            OUT                                              (        2        )            
As shown in the foregoing, prior-art voltage loop solution for boost converters have several drawbacks, in particular in terms of current sinking from a the input power supply, such as a battery, in low power applications.
For example, in order to minimize the current consumption of the feedback voltage divider, the resistors of the voltage divider should be large. However, from a practical point of view, the current IRES consumed by the voltage divider usually may not be minimized because, when increasing the resistance values of the resistors R1, R2 of the voltage divider, also their occupied area increases, rendering an integration within an integrated circuit more expansive.
On the basis of the foregoing description, the need is felt for solutions, which overcome one or more of previously outlined drawbacks.